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| Data Sheet No. PD94703 revA IRU3065(PbF) POSITIVE TO NEGATIVE DC TO DC CONTROLLER PRODUCT DATASHEET FEATURES Generate Negative Output from +5V Input 1A Maximum Output Current 1.5MHz maximum Switching Frequency Few External Components Available in 6-Pin SOT-23 DESCRIPTION The IRU3065 controller is designed to provide solutions for the applications requiring low power on board switching regulators. The IRU3065 is specifically designed for positive to negative conversion and uses few components for a simple solution. The IRU3065 operates at high switching frequency (up to 1.5MHz), resulting in smaller magnetics. The output voltage can be set by using an external resistor divider. The stability over all conditions is inherent with this architecture without any compensation. The device is available in the standard 6-Pin SOT-23. APPLICATIONS Hard Disk Drives Blue Laser for DVD R-W MR Head Bias LCD Bias GaAs FET Bias Positive-to-Negative Conversion TYPICAL APPLICATION 5V D1 BAT54 VDD Vcc C3 100pF C1 1uF C4 10uF U1 VGATE IRU3065 Gnd Q1 IRLML5203 D2 C6 10uF VOUT (-5V) L1 1.2uH R1 0.1 10BQ015 VSEN R2 10K ISEN VREF = 5V R3 10K VOUT = -VREF x R3 R2 Figure 1 - Typical application of IRU3065 for single input voltage. PACKAGE ORDER INFORMATION Basic Part (Non Lead-Free) TA (C) 0 To 70 TA (C) 0 To 70 DEVICE IRU3065CLTR DEVICE IRU3065CLTRPbF PACKAGE 6-Pin SOT-23 (L6) Lead-Free Part PACKAGE 6-Pin SOT-23 (L6) www.irf.com OUTPUT VOLTAGE Adjustable OUTPUT VOLTAGE Adjustable 1 IRU3065(PbF) ABSOLUTE MAXIMUM RATINGS Vcc ......................................................................... 7V VDD ......................................................................... 12V Operating Junction Temperature Range ..................... 0C To 125C Operating Ambient Temperature Range ..................... 0C To 70C Storage Temperature Range ...................................... -65C To +150C ESD Capability (Human Body Model) ........................ 2000V PACKAGE INFORMATION 6-PIN PLASTIC SOT-23 (L6) TOP VIEW VGATE 1 Gnd 2 VSEN 3 6 Vcc 5 VDD 4 ISEN JA=230 C/W ELECTRICAL SPECIFICATIONS Unless otherwise specified, these specifications apply over Vcc=5V, VDD=7V, CGATE=470pF, RSEN=0.125, RFDBK1=RFDBK2=10K (to Vcc), fs=1.2MHz, IFL=0.25A and TJ=0C to 125C. Typical values refer to TJ=25C. PARAMETER SYM TEST CONDITION Recommended Vcc Supply Vcc Note.1 Recommended VDD Supply VDD Operating Current Icc Initial Output Voltage Accuracy Measured in application TJ=25 C, Vout=-5V Output Accuracy Measured in application over temp. Vout=-5V. Voltage Feedback Sense VVSEN Voltage Feedback Input Offset VVoff Voltage Feedback Bias Current IVBIAS Peak Current Sense Voltage VIs Min Current Sense Voltage VIs Current Sense Bias Current IIBIAS Output Drivers Section Switching Frequency Note. 1 fs Max Output Duty Cycle Dmax Min Output Duty Cycle Dmin 10% to 90% Vgate high Rise Time Tr Fall Time Tf 90% to 10% Vgate going low Propagation Delay from TD Vsens=1V. Isens from 0 to 250mV. Delay time between Current Sense to Output 90% of Isens to 10% of Vgate MIN 4 4 -1% -2% 0 -10 145 50 2 1.5 100 0 40 40 100 10 2 TYP 5 3 1% +2% V mV A mV mV A MHz % % ns ns ns MAX UNITS V V mA Note. 1. guarantted by design 2 www.irf.com IRU3065(PbF) PIN DESCRIPTIONS PIN# 1 2 3 4 5 6 PIN SYMBOL VGATE Gnd VSEN ISEN VDD Vcc PIN DESCRIPTION Output driver for external P Channel MOSFET. This pin serves as ground pin and must be connected to the ground plane. A resistor divider from this pin to VOUT and Vcc or an external VREF, sets the output voltage. This pin sets the maximum load current by sensing the inductor current. This pin provides biasing for the output driver. This pin provides biasing for the internal blocks of the IC. BLOCK DIAGRAM Vcc 6 VDD 5 S Q R 1 VGATE 4 ISEN 3 VSEN 2 Gnd Figure 2 - Simplified block diagram of the IRU3065. www.irf.com 3 IRU3065(PbF) APPLICATION INFORMATION Introduction The IRU3065 is a controller intended for an inverting regulator solution. For example, to generate -5V from a 5V supply. The controller is simple and only has a voltage comparator, current hysteretic comparator, flipflop and MOSFET driver. It controls a typical buck boost converter configured by a P-channel MOSFET, an inductor, a diode and an output capacitor. The sensed inductor current by a sensing resistor compares with current comparator. The current comparator uses hysteresis to control the turn-on and turn-off of the transistor based upon the inductor current and gated by the output voltage level. When the inductor current rises past the hysteresis set point, the output of the current comparator goes high. The flip-flop is reset and the Pchannel MOSFET is turned off. In the mean time, the current sense reference is reduced to near zero, giving a zero reference threshold voltage level. As the inductor current passes below this threshold, which indicates that the inductor's stored energy has been transferred to the output capacitor, the current comparator output goes high and turns on the output transistor (if the output voltage is low). By means of hysteresis, the inductor charges and discharges and functions as self oscillating. The voltage feedback comparator acts as a demand governor to maintain the output voltage at the desired level. By hysteresis control, the maximum switch current (also equals inductor current) is limited by the internal current sensing reference. The power limit is automatically achieved. The switching frequency is determined by a combination of factors including the inductance, output load current level and peak inductor current. The theoretical output voltage and switching frequency versus output current is shown in Figure 3. Output voltage Regulation mode Power limit mode When the output current is below a critical current IOCP, the output voltage is regulated at the desired value and the switching frequency increases as output current increases. At current IOCP, the switching frequency reaches its maximum fS(MAX). In this region, the operation is in regulation mode. When the current goes above IOCP, the operation goes into power limit mode. The output voltage starts to decrease and the output power is limited. The switching frequency is also reduced. Analysis shows that the current IOCP is determined by: VISEN(TH) VIN IOCP = 1 x x Rs VIN-VOUT(NOM)+VD 2 --(1) Where: Rs = Current Sensing Resistance VISEN(TH) = Upper Threshold Voltage at the current comparator (when Vcc=5V, VISEN(TH)=0.145V) VIN = Input Voltage VD = Diode Forward Voltage VOUT(NOM) = Nominal Output Voltage The maximum switching frequency is determined by: fS(MAX) = fS(MAX) = VINx(VD-VOUT(NOM)) (VIN+VD-VOUT(NOM)xLxIPEAK VINx(VD-VOUT(NOM))xRS VISEN(TH)x(VIN+VD-VOUT(NOM))xL ---(2) Where: IPEAK = Peak Inductor Current IPEAK is determined by: IPEAK = VISEN(TH) RS ---(3) The detailed operation can be seen in the theoretical operation section Vout f s max Switching frequency I out fs I out I ocp Figure 3 - Theoretical output voltage and switching frequency vs. output current. 4 www.irf.com IRU3065(PbF) APPLICATION EXAMPLE Design Example The following design example is for the evaluation board application for IRU3065. The schematic is shown in figure 1: Where: VIN = 5V VOUT(NOM) = -5V IOUT = 200mA fS(MAX) = Maximum Frequency fS(MAX) = 1.2MHz VD = Diode Forward Voltage VD = 0.5V Vcc = 5V VISEN(TH)=145mV 150mV Voltage Sensing Resistor The output voltage is determined by the two voltage sensing resistors R2 and R3: VOUT(NOM) = - R3 x VREF R2 If R3 is chosen as 10K, Then R2 is given by: VREF 5V R2 = x R3 = x 10K = 10K VOUT(NOM) -5V Current Sensing Resistor RS In order to select RS, the desired critical current IOCP has to be determined. Considering the switching losses, for conservative, the critical current should select to be slightly greater than the nominal output current. Select: IOCP = 200mAx1.5 = 300mA Where 1.5 is the coefficient to take the efficiency into account. According to equation (1), the current IOCP is given by: IOCP = VIN 1 0.15 x x = 300mA RS VIN - VOUT(NOM) + VD 2 VIN 1 0.15 x x 2 IOCP VIN - VOUT(NOM) + VD The modified current IOCP is: 0.15 VIN 1 x x RS VIN + VD - VOUT(NOM) 2 5 1 IOCP = x x 1.5A = 357mA 5 + 0.5 - (-5) 2 IOCP = Output Inductor L The inductance is chosen by equation (2): L L VINx(VD - VOUT(NOM)) (VIN+VDVOUT(NOM))xfS(MAX)xIPEAK -(-5 - 0.5) 5 x (5 - (-5) + 0.5)x1.2MHz 1.5A = 1.45H Select L = 1.2H The maximum inductor current is: IPEAK = 1.5A The maximum average inductor current equals IAVG=(VISENTH_MAX+VISENTH_MIN)/Rs/2 IAVG=(145mV+50mV)/0.1ohm/2=1A MOSFET Selection A P-channel MOSFET is required. The peak current in this case is equal to IPEAK=1.5A. The MOSFET IRLML5203, from international Rectifier with ID=3A and BVDSS=30V, is a good choice. Input Capacitor An input capacitor will help to minimize the induced ripple on the +5V supply. A 1F to 10F X7R ceramic capacitor is recommended. Output Capacitor An output capacitor is required to store energy from transfer to the output inductor. Its capacitance and ESR have a great impact on output voltage ripple. A 10F to 22F X7R Tantolum or ceramic capacitor is recommended. Output Diode The average diode current equals output current. In this case, select the diode average current larger than 300mA. The lowest block voltage is VIN+(-VOUT). In this case, It is 10V. In order to reduce the switching losses, the Schottky diode is recommended. The diode 10BQ015 from International Rectifier with ID=1A and VBR=15V is a good choice. Other Components In order to speed up the turn off of P-channel MOSFET, a fast diode 1N4148 or a 100ohm resistor and 100pF capacitor is connected to the pin VDD and VGATE as shown The current sensing resistance is calculated as: RS = RS = 5 1 0.15 x x y 0.12 0.3 5 - (-5) + 0.5 2 Select RS = 0.1 From equation (3), the modified inductor peak current is: VISEN(TH) IPEAK = = 1.5A RS www.irf.com 5 IRU3065(PbF) in figure 1. The schottky diode can be replaced with a 100 resistor (Figure 28.) with a small sacrifice of efficiency but lower cost. Thermal Consideration The thermal design is to ensure maximum junction temperature of IRU3065 will not exceed the maximum operation junction temperature, which is 125 C. The junction temperature can be estimated by the following: TJ = PDxJA+TATJ(MAX) = 125 C Where JA is the thermal resistance from junction to case which is usually provided in the specification. PD is the power dissipation. TA is the ambient temperature. The package thermal resistance of IRU3065 is estimated as 230C/W due to compact package. Assuming the maximum allowed ambient temperature is 70C, the maximum power dissipation of IRU3065 will be PD<(125-70C)/JA=(125-70)/230=240mW For High Power Application The IR3065 driver is designed to driver PMOS for low current applications. Figure 4. shows the rise time versus cap load. For big capacitor load, the rise time is increasing. The measured frequency versus load is listed in Figure 8. The highest switching frequency occurs at about 440mA. As load current goes up, the IC goes into power limit mode and frequency automatically goes down to protect the system. The current sensing comparator threshold voltage versus VCC is shown in Figure. 9. Since this threshold is only a divided voltage of VCC, it will changes when VCC changes. This should be aware in the application. The output voltage versus Vin=VCC is shown in figure 11. Since the voltage reference is set by Vin. When Vin changes, the output voltage will change along Vin. Sometimes this feature is preferrable since Vout may want to be tracked with Vin except the polarity. However, if more accurate output is required, a external voltage reference should set the output voltage. For the evaluation board, the measured inductor voltage waveforms are listed in Figure 13-17. Figure 15 shows the measured inductor voltage waveform when output current is 250mA, which the converter is operated in regulation mode and output voltage is regulated at desired voltage -5V. Figure 16 shows the measured inductor voltage waveform when the output current is equal to the critical current IOCP. Figure 17 shows the measured inductor voltage waveform when the output is in short circuit, which indicates that the converter is in power limit mode and output voltage is near zero. Demo board Evaluation Results Fig.1 shows the evaluation board schematic and the selected components. The diode D1 can be replaced with a 100ohm resistor. The measured efficiency versus load current is shown in Fig. 6. With the boot strap schottky diode, the efficiency is slight higher comparing with using 100ohm resistor. If higher efficiency is preferred, lower operation frequency should be selected. Figure. 5 shows a efficiency curve when 4.7uH inductor is chosen. The maximum operation frequency reduces from 800k to 250kHz. As a results, efficiency is more than 10% higher. For the application circuit shown in Fig.1. The measured output voltage versus output current is shown in Figure 7. When the load current approaches 400mA, the output voltage starts to drop and goes into power limit mode. When output is about 1A, the output voltage will goes almost zero. Rise time versus cap load 90 80 70 rise time(ns) 60 50 40 30 20 10 0 0 0.5 1 1.5 2 2.5 Cap(nF) Fig.4. Rise time versus cap load. The internal gate driver of IRU3065 is designed for load current up to 1A. For higher power applications, external driver is recommended to driver the external FETs. 6 . www.irf.com IRU3065(PbF) Characteristics of IRU3065 Efficiency versus load current 6 Vout(V) vs Iout(mA) 75 Efficiency(%) 70 60 55 50 0 100 200 Iout(mA) 5 4 Vout(V) 3 2 65 300 400 1 0 0 200 400 600 800 1000 Iout(mA) Efficiency(%) with diode Efficiency(%) with 100ohm resistor Figure.5 Efficiency with 4.7uH inductor, 250kHz operation Fig.7. Output voltage (absolute value) versus load current. (Vout= -5V, Iocp=400mA) Efficiency versus load current Frequency (KHz) versus load current 75 Efficiency(%) 70 65 60 55 50 0 100 200 300 400 Iout(mA) Efficiency(%) with diode Efficiency(%) with 100 resistor Frequency(kHz) 900 800 700 600 500 400 300 200 100 0 0 200 400 600 800 1000 Iout(mA) Figure 6. Efficiency with 1.2uH inductor, 800k Hz operation Fig.8. Frequency versus load current. (Vout= -5V, Iocp=400mA). www.irf.com 7 IRU3065(PbF) Characteristics of IRU3065( Continued) Current comparator threshold versus Vin (T A =25C) 180 170 Isen(th) MV 160 150 140 130 120 4.5 4.7 4.9 Vin 5.1 5.3 5.5 Output voltage versus Vin@Iout=200mA, TA=25C -4 -4.2 4.5 -4.4 Output voltage(V) -4.6 -4.8 -5 -5.2 -5.4 -5.6 -5.8 -6 VIn 4.7 4.9 5.1 5.3 5.5 Figure 9. Current sensing comparator upper threshold versus VCC=Vin. Figure. 11. Output voltage versus Vin (Vcc=Vin). Current sensing comparator upper threshold (mV) versus temperature 155 154 153 151 150 149 148 147 146 145 144 -25 0 25 50 75 100 125 Temperature(C ) Output voltage(V) -5 Output voltage versus temperature (Vin=5V,Iout=200mA) -25 -5.02 -5.04 -5.06 -5.08 -5.1 -5.12 -5.14 0 25 50 75 100 125 152 Tempeture(C ) Figure. 10. Current sesning comparator upper threshold versus temperature (Vcc=5V) Figure. 12. Output voltage versus temperature at Vcc=Vin=5V and Iout=200mA. 8 www.irf.com IRU3065(PbF) . Operation Waveforms of Demo board in Figure. 5 Figure. 13 Start up Fig. 16. Operation waveform with 450mA, the boundary of continuous mode and discontinuous mode. The output start out of regulation Fig. 14. Operation waveform with 20mA load. Fig. 17. operation waveform with short output. Fig. 15. Operation waveforms with 250mA load (normal operation) www.irf.com 9 IRU3065(PbF) THEORETICAL OPERATION Operation-Regulation Mode reference of the chip, which is set to be 150mV (for Vcc=5V), the flip-flop is reset and the PMOS is turned off. The inductor current is discharged through diode D2 to the load. The load voltage increases. When the inductor current decreases to zero, the output current is supplied by the output capacitor and the output voltage decreases until next cycle starts. In this mode, the voltage at VSEN pin is controlled near zero. Therefore, the output voltage is regulated at: R3 -VOUT = x VREF R2 In the evaluation board, the output voltage is regulated at -5V, as shown in figure 7. The steady state of the converter should be operated in this mode. One feature in this mode is that the shaded inductor current in figure 18 stays unchanged. The average output diode current equals output current. When the switching period decreases and frequency goes up, the average diode current increases to support more output current. The switching frequency increases linearly when the load current increases as shown in figure 20. 6 5 4 V g a te Vi n V o l g e a cro ss ta th e i d u cto r n V out - V D In d u cto r cu rre n t Ipeak O u tp u t o f cu rre n t co m p a ra to r O u tp u t d i d e o cu rre n t Iout V out I out 5 - V out R3 = V ref R2 ton Ts t1 3 2 1 0.75 0 0 0.02 0.16 0.32 I out 0.48 0.64 0.8 0.8 Figure 19 - Theoretical output voltage (-VOUT) versus output current for IRU3065 controlled buck boost evaluation board.(assume VIsen=0.2V) 1.083 .10 1.5 .10 6 1.2 .10 9 .10 f s I out 6 .10 3 .10 4.583 .10 4 6 Figure 18 - Operation waveforms of IRU3065 controlled buck boost converter at regulation mode. In general, IRU3065 controlled buck boost converter is operated in two modes depending on the load current. When the load current is small, the buck boost operated in first mode (regulation mode). The operation waveforms are shown in figure 18. In this mode, the inductor current in the buck converter is discontinuous. Basic Operation When the voltage at VSEN pin is below zero, the flip-flop inside the IC is set and the VGATE pin output low, which trigger the PMOS in the power stage, the output inductor current increases from zero. When sensed inductor current voltage at ISEN pin reaches the internal current 6 5 5 5 0 0.02 0 0.2 0.4 I out 0.6 0.8 0.8 Figure 20 - Theoretical switching frequency versus output current for evaluation board.(assume VIsen=0.2V) 10 www.irf.com IRU3065(PbF) Power Limit Mode When the output current continuous increases, the switching period continuous decreases until the inductor current goes into the boundary of discontinuous and continuous mode as shown in Figure 21. Then the IRU3065 controlled buck boost converter goes into power limit mode. In this mode, the output power is limited. The output voltage is no longer regulated. The output voltage decreases when the load current increases as shown in Figure 19. In this mode, the shaded inductor current in Figure 18 keeps same. The turn off time period is dependent on the output voltage. When the output current increases, the output voltage decreases and it takes more time for the inductor current to reset from peak current to zero. Therefore, the turn off period increases. Overall the switching frequency decreases when load current increases as shown in Figure 20. Influence of System Parameters From above section, there is a critical output current IOCP. When the output current is larger than IOCP, the output voltage is out of the regulation and switching frequency starts to decreases. When output current equals IOCP, the frequency reaches its maximum fS(MAX). Analysis shows that the current IOCP and maximum frequency fS(MAX) strongly depends on the parameters such as current sensing resistor RS and inductance L as well as the input and output voltage. 5 6 5 V out I out , 0.1 . V out I out , 0.11 . V out I out , 0.12 . 4 3 2 1 Vgate 0.452 0 0 0.02 0.1 0.2 0.3 I out 0.4 0.5 0.6 0.7 0.7 Figure 22 - Theoretical output voltage versus output current with different current sensing resistor RS. Vin Voltage across the inductor 6 1.5 .10 6 1.3 .10 6 1.2 .10 f s I out , 1 .H f s I out , 1.1 .H f s I out , 1.2 .H 5 9 .10 5 6 .10 5 3 .10 Vout - VD Inductor current I peak 4 4.583 .10 0 0.02 0 0.1 0.2 0.3 I out 0.4 0.5 0.6 0.7 0.7 Output of current comparator Output diode current Figure 23 - Theoretical operation switching frequency versus output current with different inductance L. Figure 22 shows the calculated output voltage versus output current with different current sensing resistor RS. With different RS, the critical current IOCP varies, and the power process ability changes. Figure 23 shows the calculated operation switching frequency versus output current with different inductance L when RS=0.1. The inductance L determines the maximum switching frequency of the buck boost converter. I out R3 Vref R2 - Vout t on t1 Ts Figure 21 - Operation waveforms of IRU3065 controlled buck boost converter at power limit mode. www.irf.com 11 IRU3065(PbF) Analysis of Operation Regulation Mode From Figure 18, when the PMOS is on, the inductor current increases from zero. That is: IL = VIN xt L ---(4) The expected switching frequency linearly increases as output current goes up, as shown in Figure 20. Power Limit Mode When output current continuously increases and IOUT=IOCP, the converter is in the boundary of regulation mode and power limit mode with output voltage is regulated to nominal voltage VOUT=VOUT(NOM). As current continues to increase (IOUT>IOCP), the converter goes into power limit mode. In this mode, the maximum inductor current is limited by the internal current reference VISEN=145mV. Therefore, the turn on time of the PMOS keeps same as equation (7). For turn off time, the inductor current theorectically should decrease from IPEAK to zero if the threshold voltage is close to zero , therefore: t1 = L x IPEAK VISEN x L = -(VOUT - VD) -(VOUT - VD) x RS ---(12) And the peak current is given by: IPEAK = VIN x tON L ---(5) Where tON is the turn on time of the PMOS. Because the switch is turned off when sensed inductor current reaches threshold VISEN, the following equation holds: VIN xtON = VISEN=150mV ---(6) RSxIPEAK = RSx L VISEN(TH) IPEAK = RS The turn on time of the PMOS can be calculated as: LxIPEAK VISENxL tON = = VIN RSxVIN ---(7) Where VD is the forward voltage drop of output diode D2. The switching period is given by: TS = tON + t1 = L x IPEAK L x IPEAK + VIN -(VOUT - VD) ---(13) For inductor, by applying voltage and second balance approach, we have: VINxtON+(VOUT - VD)xt1 = 0 It can be derived as: VINxtON VISENxL t1 = -(VOUT - VD) = -(VOUT - VD)xRS ---(8) VIN - VOUT + VD TS = L x IPEAK x -VIN x(VOUT - VD) Where VD is the forward voltage drop of output diode D2. From Figure 18, the average current of output diode should equals the output current, resulting in: ID(AVG) = 1 t1 x IPEAK x = IOUT 2 TS ---(9) The combination of equations (12) and (13) result in the following: VIN t1 = TS VIN - VOUT + VD ---(14) The output current equals the average diode current, which is: IOUT = IOUT = 1 t1 xIPEAKx 2 TS 1 VISEN VIN x x 2 RS VIN - VOUT + VD ---(15) 1 Where TS is the switching period and fS = TS Combination of equation (6)(8)(9) results in the relationship between output current and switching frequency: fS = -RS2x(VOUT - VD) xIOUTx2 VISENxVISENxL ---(10) Where the peak current is given by equation (6). Equation (15) can be rewritten as: VOUT = VIN + VD VISEN x VIN 2RS x IOUT ---(16) Because at regulation mode, the output voltage is regulated, i.e. VOUT=VOUT(NOM). Then the equation (10) can be rewritten as: fS = -RS2x(VOUT(NOM) - VD) xIOUTx2 VISENxVISENxL ---(11) The above equation shows that the output voltage at the power limit mode is not regulated. It decreases as the output current increases. 12 www.irf.com IRU3065(PbF) When IOUT=IOCP, the output voltage equals nominal voltage VOUT=VOUT(NOM). From equation (15),we have IOCP = 1 VISEN VIN x x 2 VIN - VOUT +VD RS ---(17) 6 Vout versus output current Output voltage The above equation is used to select the current sensing resistor RS. Substitution of equation (16) into equation (13) results in the relationship between frequency and output current, that is VIN 2xIOUT fS = x 1LxIPEAK IPEAK 5 4 3 2 1 0 0 0.2 0.4 0.6 0.8 ( )--(18) Output current (A) Predicted (-Vout) Measured -Vout The above equation indicates that the switching frequency decreases when output current increases during power limit mode. When IOUT=IOCP, the switching frequency reaches its maximum. Substitution of VOUT=VOUT(NOM) and equation (6) into equation (13) results in the maximum switching frequency: VINx(VD - VOUT(NOM)) fS(MAX) = (VIN + VD VOUT(NOM))xLxIPEAK VINx(VD - VOUT(NOM))xRS fS(MAX) = ---(19) VISENx(VIN + VD VOUT(NOM))xL Therefore, the inductance can be selected according to the maximum desired frequency as shown in the following: VINx(VD - VOUT(NOM)) (VIN + VD VOUT(NOM))xfS(MAX)xIPEAK Figure 24- The comparison between predicted and measured output voltage versus output current Switching frequency versus output current Frequency(KHz) 1200 1000 800 600 400 200 0 0 0.2 0.4 0.6 0.8 Output current (amp) Predicted fs(KHz) Experiment fs (kHz) L ---(20) Fig. 24 and Fig.25 shows the theorectical predication and calculation results for the output voltage and frequency versus output current. Figure 25 - The comparison between predicted and measured switching frequency versus output current www.irf.com 13 IRU3065(PbF) Other Applications 5V 100ohm VDD Vcc C1 100pF U1 VGATE IRU3065 Gnd Q1 IRLML5203 L1 1.2uH R1 0.1 D2 C2 10uF 10BQ015 VOUT (-5V) VSEN ISEN VREF= 5V R2 10K R3 10K Fig. 26 . IRU3065 application with 100ohm resistor and 100pf cap 14 www.irf.com IRU3065(PbF) (L6) SOT-23 Package B e L E E1 e1 D C A2 A C L A1 SYMBOL A A1 A2 B C D E E1 e e1 L MAX MIN 1.45 0.90 0.15 0.00 1.30 0.90 0.50 0.35 0.20 0.09 3.00 2.80 3.00 2.60 1.75 1.50 0.95 REF 1.90 REF 0.60 0.10 10 0 NOTE: ALL MEASUREMENTS ARE IN MILLIMETERS. IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 9/6/2005 www.irf.com 15 |
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